1. Field of the Invention
The present invention relates to a self-oscillation type switching power supply, and more particular to a switching power supply outputting a high voltage.
2. Description of the Related Art
Ringing-choke converters have been used as self-oscillation types switching power supplies. FIG. 2 is a circuit diagram of a conventional ringing choke converter. In the diagram, numeral 11 shows a direct current (hereinafter, DC) power circuit that rectifies and smoothes a commercial alternating-current (hereinafter, AC) power AC and generates a DC voltage of about 120 V, and symbol T is a transformer having a primary winding Lp, a secondary winding Ls, and a feedback winding Lf. Symbol Q1 shows a switching transistor that is connected to the DC power supply via the primary winding Lp of the transformer. A starting resistor R1 is connected to the base of this switching transistor Q1. A current limiting resistor R2, an accelerating capacitor C2, and a diode D2 are connected between the feedback winding Lf and base of the switching transistor Q1. In addition, a control transistor Q2 is connected between the base and emitter of the switching transistor Q1, and a time-constant circuit 4 comprising a resistor 5 and a capacitor C3 is provided in the feedback winding Lf. The time-constant circuit 4 is connected so that a voltage of the capacitor C3 may be applied to the base of the control transistor Q2. A rectifying and smoothing circuit 2 comprising a rectifier diode D1 and a smoothing capacitor C1 is connected to the secondary winding Ls of the transformer T. A resistor voltage-dividing circuit comprising resistors R3 and R4, a current limiting resistor R10, a variable shunt regulator 12, and a light emitting diode of a photocoupler PC are connected to the output side of this rectifying and smoothing circuit 2. A phototransistor of this photocoupler is connected to a charging path of the capacitor C3.
Operation of a power supply apparatus shown in FIG. 2 is as follows. When a DC voltage is applied from the DC power circuit 11, a minute starting current flows to the base of the switching transistor Q1 via the starting resistor R1. Owing to this, if a voltage between the collector and emitter between the switching transistor Q1 is decreased since a current flows in the collector of the transistor Q1, a voltage is applied between terminals of the primary winding Lp of the transformer T, and an induced voltage proportional to this voltage is generated in the feedback winding Lf. Since a positive feedback current is supplied to the base of the switching transistor Q1 via the current limiting resistor R2, accelerating capacitor C2, and diode D2, the transistor Q1 turns ON (saturated). If the transistor Q1 turns ON, a DC voltage is applied between terminals of the primary winding Lp of the transformer T, and a current flows in the primary winding Lp to excite the transformer T. At this time, an induced voltage generated simultaneously in the feedback winding Lf charges the capacitor C3 via the resistor R5 and accelerating capacitor C2, diode D2, and phototransistor of the photocoupler PC. If the charged voltage of the capacitor C3 reaches a threshold voltage (about 0.6 V) between the base and emitter of the control transistor Q2, the base and emitter of the switching transistor Q1 are short-circuited, and hence a base current of the switching transistor Q1 is cut off to cut out the transistor Q1 sharply. If the switching transistor Q1 goes OFF, the induced voltage of the feedback winding Lf reversely biases the base of the switching transistor Q1 to a negative potential. At the same time, the feedback winding Lf discharges the capacitor C3 via the resistor R5, and hence the base of the control transistor Q2 is reverse-biased at a negative potential. Therefore, an OFF period is continued until all the excited energy of the transformer T is released from the secondary winding Ls. If all the excited energy of the transformer T is released, the induced voltage of the feedback winding Lf abruptly disappears, but a ringing voltage (kick voltage) is generated in the direction where the base of the switching transistor Q1 is forward-biased by leakage inductance and distributed capacitance of the transformer T to turn on the switching transistor Q1 again. After that, oscillation grows and continues with repeating ON/OFF operation described above.
Here, let a voltage between both ends of the rectifying and smoothing circuit 2 be an output voltage Vout, let a current passing a load be Iout, let inductance of the primary winding Lp be Lp, and let the peak value of a collector current of the switching transistor Q1 be Icp, and the output voltage Vout can be approximated by the following formula: EQU Vout=(Lp.multidot.Icp.sup.2)/(2Iout) (1)
In addition, let ON time of the switching transistor Q1 be ton, and let a voltage applied between terminals of the primary winding Lp be Vin, and the current Icp can be expressed by the following formula: EQU Icp=(Vin/Lp)ton (2)
According to the relation expressed in formulas (1) and (2), it is possible to maintain the output voltage Vout to be constant by adjusting the current in the phototransistor of the photocoupler PC through detecting the output voltage and by controlling the ON time ton of the switching transistor Q1.
Nevertheless, in a conventional self-oscillation type switching power supply shown in FIG. 2, the output voltage Vout is a low voltage such as 5 V, and the transformer T is a step-down transformer. Although it becomes possible to configure at once a power supply apparatus generating a high voltage by increasing a turn ratio of the secondary winding Ls to the primary winding of the transformer T in the configuration of the conventional power supply apparatus shown in FIG. 2, problems arise, as discussed below.
FIG. 3 is a circuit diagram of a transformer, symbol Cs shows distributed capacitance generated between terminals of the secondary winding Ls, and Cps is other distributed capacitance generated between the primary winding Lp and secondary winding Ls. In addition, symbol Cpp shows capacitance into which the distributed capacitance Cs and Cps is converted as capacitance between terminals of the primary winding Lp. Although a power supply apparatus boosting a DC voltage input of some tens of volts to a DC or AC voltage of some hundreds through some thousands of volts is requested in, for example, an electrophotography type of copy machine or page printer, so as to obtain such a characteristic, it is necessary to extremely increase the turn ratio of the secondary winding Ls to the primary winding Lp in a high-voltage transformer. Here, let the number of turns of the primary winding Lp be Np, let the number of turns of the secondary winding Ls be Ns, and let values of the distributed capacitance Cs and Cps be Cs and Cps respectively, and the distributed capacitance Cpp into which the capacitance Cs and Cps are converted as the capacitance between terminals of the primary winding Lp can be approximated by the following formula: EQU Cpp=(Cs+Cps).times.(Ns/Np).sup.2 (3)
Therefore, in the high-voltage transformer, the capacitance Cpp becomes an excessively large value in comparison with the low voltage transformer. In addition, let inductance of the primary winding Lp be Lp, and an inherent parallel resonance frequency fo configured with the inductance Lp of the primary winding and the primary side-converted capacitance Cpp is expressed by the following formula: EQU fo=1/(2.pi.(Lp.multidot.Cpp).sup.1/2) (4)
Based on this formula, the above-described resonance frequency fo in the high-voltage transformer is a lower frequency in comparison with that in the low voltage transformer.
Then, a point largely different from the above-described operation in the case that the transformer in FIG. 2 is replaced with the high-voltage transformer shown in FIG. 3 is that the high-voltage transformer freely oscillates at the resonance frequency fo determined by formula (4) during a period from the switching transistor Q1 turning off to when it next turns off. In the conventional low voltage switching power supply shown in FIG. 2, an oscillation frequency largely changes according to output power consumption. For example, as the output power consumption becomes small, the low voltage transformer T can be excited in further minute ON time, and in consequence, the oscillation frequency is apt to increase. Since the low voltage transformer T has excessively small capacitance Cpp shown in formula (3) and a high inherent resonance frequency, the low voltage transformer T can oscillate in the frequency range of some hundreds of kHz, but, in the high-voltage transformer, it becomes difficult to oscillate at a frequency higher than the inherent frequency even in the case of the output power consumption being in the state of no load since the inherent resonance frequency fo, as described above, is very low.
FIG. 4 is an equivalent circuit of a circuit of the high-voltage transformer shown in FIG. 3 and the switching transistor. Here, symbols L1 and L2 show leakage inductance, Lp is the excited inductance of the primary winding, and Cpp is the primary side-converted distributed capacitance shown in FIG. 12. Here, let an inductance component of the leakage inductance L1 and L2 be L.sub.1e, and a serial resonance frequency fo' is expressed by the following formula: EQU fo'=1/(2.pi..sqroot.(L.sub.1e .multidot.Cpp)) (5)
As described above, since the primary side-converted capacitance Cpp is extremely large in the high-voltage transformer, the serial resonance frequency fo' shown in formula (5) also becomes comparatively low. Although the serial resonance frequency fo' depends on a value of the leakage inductance L.sub.1e, as an order, the resonance frequency fo' becomes a frequency component that is generally about 6-10 times the parallel resonance frequency expressed by formula (4), that is, near to it. Therefore, if such a high-voltage transformer is applied to the circuit shown in FIG. 2, a ringing component is superimposed on a voltage Vce between the collector and emitter of the switching transistor Q1 , as shown in FIGS. 5A to 5C.
In addition, although a wide range of variable performance on an output voltage (current) is requested in a high voltage power supply apparatus, if the ON time of the switching transistor Q1 is adjusted so as to change in a wide range a voltage induced between the secondary winding Ls of the high-voltage transformer, the OFF time of the switching transistor Q1, as described above, is determined by the parallel resonance frequency fo, and hence toff.apprxeq.1/(2fo), that is, toff is constant. On the other hand, although the frequency of a ringing component is fo' that is shown in formula (5) and is constant, the leakage inductance L.sub.1e, is an extremely small value in comparison with the inductance Lp of the primary winding, and hence, even if the ON time of the switching transistor Q1 is changed, the amplitude of the ringing component is changed to not so large an extent. Therefore, as the ON time of the switching transistor Q1 becomes short since the voltage between the collector and emitter of the switching transistor Q1 is changed as shown in FIGS. 5A to 5C, the serial resonance frequency component shown in formula (5) becomes dominant.
According to the conventional circuit shown in FIG. 2, its construction is that a voltage induced in the feedback winding Lf discharges the capacitor C3 and a voltage induced in the feedback winding Lf and a current of the phototransistor of the photocoupler PC charges the capacitor C3, and hence, if the ON time ton of the switching transistor Q1, as shown in FIG. 5C, becomes near to the period 1/(2fo') to which the frequency of the ringing component is determined, the time ton follows the serial resonance frequency fo' since the time constant circuit 4 comprises passive elements, and in consequence, the control transistor Q2 also follows the serial resonance frequency for the switching transistor Q1 to reach the serial resonance operation, not the parallel oscillation in which the switching transistor Q1 should operate. Owing to this, the relation between the ON time of the switching transistor Q1 and the output voltage Vout becomes non-linear, circuit operation becomes unstable like intermittent oscillation, and hence stable control cannot be performed. Furthermore, since switching loss increases by the switching transistor Q1 turning on/off at a high frequency, upsizing of a heat sink becomes necessary.
In order to solve the above-described problems, the ON time of the switching transistor Q1, as shown in FIG. 6, is not adjusted in a conventional high voltage power supply apparatus, but it is common that the DC input voltage inputted to the primary winding of the high-voltage transformer is adjusted to stabilize its output. In FIG. 6, symbol Q5 shows a control power transistor for performing step-down of a voltage of a DC input power supply 1 and thereby adjusting an input voltage to the high-voltage transformer. A control circuit controls a base current of the transistor Q5 according to a detection signal from an output voltage detection circuit to stabilize an output voltage. The switching transistor Q1 always performs ON/OFF operation in a constant period by an oscillation circuit.
Nevertheless, since the circuit shown in FIG. 6 is configured in a separately-excited oscillation type switching power supply, an external oscillator becomes necessary. Further, another power transistor for performing step-down of the input voltage to the high-voltage transformer becomes necessary, and hence the circuit configuration becomes complicated and large.